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Gate level dynamic energy estimation in asynchronous circuits using Petri nets

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Gate level dynamic energy estimation in asynchronous circuits using Petri nets
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Mabry, Ryan
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Simulation
Power
Handshaking
Modeling
CPN tools
Dissertations, Academic -- Computer Engineering -- Masters -- USF   ( lcsh )
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theses   ( marcgt )
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Abstract:
ABSTRACT: This thesis introduces a new methodology for energy estimation in asynchronous circuits. Unlike existing probabilistic methods, this is the first simulative work for energy estimation in all types of asynchronous circuits. The new simulative methodology is based on Petri net modeling. A real delay model is incorporated to capture both gate delays and interconnect delays. The switching activity at each gate is captured to measure the average dynamic energy consumed per request/acknowledge handshaking pair. The new type of Petri net is called Hierarchical Colored Asynchronous Hardware Petri net (HCAHPN). The HCAHPN is able to capture the temporal and spatial correlations of signals within a circuit, while preserving gate logic behavior and timing information. While Petri nets have been previously used for simulating combinational and sequential circuits, this is the first work that uses Petri nets for simulating asynchronous circuits. While different asynchronous design styles make various assumptions on the gate and wire delays present with the circuit, the physical implementations of these circuits always have gate and interconnect delays. Unlike previous methods, the proposed methodology is independent of the asynchronous design style used and it can be adapted for all types of asynchronous circuits that use handshaking communication.
Thesis:
Thesis (M.S.)--University of South Florida, 2007.
Bibliography:
Includes bibliographical references.
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Statement of Responsibility:
by Ryan Mabry.
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Document formatted into pages; contains 78 pages.

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Gate Level Dynamic Energy Estimation In Asynchronous Circuits Using Petri Nets by Ryan Mabry A thesis submitted in partial fulfillment of the requirements for the degree of Master of Science in Computer Engineering Department of Computer Science and Engineering College of Engineering University of South Florida Co-Major Professor: Nagarajan Ranganathan, Ph.D. Co-Major Professor: Hao Zheng, Ph.D. Srinivas Katkoori, Ph.D. Date of Approval: June 20, 2007 Keywords: Simulation, Power, Handshaking, Modeling, CPN Tools Copyright 2007, Ryan Mabry

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ACKNOWLEDGEMENTS I would like to thank Dr. Ranganathan and Dr. Zheng for directing and helping me throughout this thesis. I am also thankful for the help provided by Narender Hanchate, Ashok Murugavel, and Soumyaroop Roy. I would also like to thank my family for the support and encouragement they have shown. On the professional level, I would like to thank the many institutions and people that developed the tools that made this thesis possible. I would like to thank the University of Manchester and Dr. Steve Furber for the development of their Balsa asynchronous synthesis tool. I would also like to thank the Virginia Tech VLSI for Telecommunications group and Dr. Dong Ha for the development of their cell library. I would also like to thank Dr. Roberto Tamassia and Brown University for the development of the JDSL library, which this work made extensive use of. I would also like to thank the CPN group at the University of Aarhus in Denmark for their development of CPN Tools.

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i TABLE OF CONTENTS LIST OF TABLES iii LIST OF FIGURES iv ABSTRACT vi CHAPTER 1 INTRODUCTION 1 1.1 The Need for Low Power 1 1.2 Power Consumption 3 1.2.1 Static Power Consumption 3 1.2.2 Dynamic Power Consumption 4 1.3 Asynchronous Circuits 5 1.3.1 Asynchronous Communication 5 1.3.2 Asynchronous Circuit Implementation Styles 7 1.4 Motivation 7 1.5 Contributions 8 1.6 Related Work 9 1.6.1 Sequential and Combinational Power Estimation 10 1.6.2 Asynchronous Energy and Power Estimation 15 1.7 Thesis Overview 19 CHAPTER 2 PETRI NET FRAMEWORK 20 2.1 Petri Net Basics 20 2.2 Power Estimation Using Petri Nets 25 CHAPTER 3 PETRI NET MODELING OF ASYNCHRONOUS CIRCUITS FOR ENERGY ESTIMATION 29 3.1 Hierarchical Colored Asynchronous Hardware Petri Nets 29 3.2 HCAHPN Low-Level Modeling 33 3.2.1 Gate Functionality Substructure 34 3.2.1.1 Gate Functionality Firing Example 38 3.2.2 Switching Activity Substructure 45 3.2.2.1 Switching Activity Firing Example 47 3.2.3 Token Filter Substructure 51 3.2.3.1 Token Filter Firing Example 53 3.3 HCAHPN High-Level Modeling 56

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ii CHAPTER 4 A FRAMEWORK FO R ENERGY ESTIMATION 61 4.1 Tool Flow 61 4.2 Energy Estimation Formula 66 CHAPTER 5 EXPERIMENTAL RESULTS 68 CHAPTER 6 CONCLUSION 71 REFERENCES 73

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iii LIST OF TABLES Table 1.1 ITRS Road-Map, 2006 International Technology Road-Map for Semiconductors 2 Table 4.1 Intrinsic Gate Delays 64 Table 5.1 Simulation Results 68

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iv LIST OF FIGURES Figure 1.1 As ynchronous Communication 5 Figure 1.2 Two-Phase Handshaking 6 Figure 1.3 Four-Phase Handshaking 6 Figure 1.4 Taxonomy Diagram of Power Estimation at Different Levels of Abstraction in Sequential and Combinational Systems 10 Figure 1.5 Taxonomy Diagram of Energy and Power Estimation Methods in Asynchronous Systems 15 Figure 1.6 Petri Net Repr esentation of Muller C-element 17 Figure 2.1 HCHPN St ructure of NAND Gate 26 Figure 2.2 Tool Flow for HCHPN Power Estimation 28 Figure 3.1 AND Gate 33 Figure 3.2 HCAHPN Structure of AND Gate 33 Figure 3.3 Gate Functiona lity Structure Detail of AND Gate 34 Figure 3.4 Gate Functionality Step 0 38 Figure 3.5 Gate Functionality Step 1 39 Figure 3.6 Gate Functionality Step 2 40 Figure 3.7 Gate Functionality Step 3 41 Figure 3.8 Gate Functionality Step 4 42 Figure 3.9 Gate Functionality Step 5 43 Figure 3.10 Gate Functionality Step 6 44

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v Figure 3.11 Switching Activit y Structure Detail of AND Gate 45 Figure 3.12 Switching Activity Step 0 47 Figure 3.13 Switching Activity Step 1 48 Figure 3.14 Switching Activity Step 2 49 Figure 3.15 Switching Activity Step 3 50 Figure 3.16 Token Filter Structure Detail of AND Gate 51 Figure 3.17 Unknown Signal State 52 Figure 3.18 Known Signal State 52 Figure 3.19 Token Filter Step 0 53 Figure 3.20 Token Filter Step 1 54 Figure 3.21 Token Filter Step 2 55 Figure 3.22 Wine Shop Asynchronous Circuit 56 Figure 3.23 Flatte ned Wine Shop Circuit 57 Figure 3.24 Wine Shop Gate Signal Graph 58 Figure 3.25 Wine Shop Petri Net 59 Figure 3.26 Wine Shop HCAHPN 60 Figure 4.1 Proposed Fr amework for Energy Estimation 61 Figure 4.2 Verilog Netlist Flattener 62

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vi Gate Level Dynamic Energy Estimation In Asynchronous Circuits Using Petri Nets Ryan Mabry ABSTRACT This thesis introduces a new methodology for energy estimation in asynchronous circuits. Unlike existing probabilistic methods, this is the first simulative work for energy estimation in all types of asynchronous circuits. The new simulative methodology is based on Petri net modeling. A real delay model is incorporated to capture both gate delays and interconnect delays. The switching activity at each gate is captured to measure the average dynamic energy consumed per request/acknowledge handshaking pair. The new type of Petri net is called Hierarchical Colored Asynchronous Hardware Petri net (HCAHPN). The HCAHPN is able to capture the temporal and spatial correlations of signals within a circuit, while preserving gate logic behavior and timing information. While Petri nets have been previously used for simulating combinational and sequential circuits, this is the first work that uses Petri nets for simulating asynchronous circuits. While different asynchronous design styles make various assumptions on the gate and wire delays present with the circuit, the physical implementations of these circuits always have gate and interconnect delays. Unlike previous methods, the proposed methodology is independent of the asynchronous

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vii design style used and it can be adapted for all types of asynchronous circuits that use handshaking communication.

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1 CHAPTER 1 INTRODUCTION Power estimation is a crucial part of the ASIC design flow. If the estimated amount of power consumed by a circuit is too low, it could exceed its operating parameters and overheat; if it is too high, companies may spend unnecessary time and money on heat reduction equipment. Asynchronous circuits are circuits that have no global clock. With the absence of a global clock, the only parts of an asynchronous circuit that will be contributing to energy consumption are those that have been requested to perform a processing action. Due to this functionality, asynchronous circuits are inherently low-powered. In this chapter, an introduction to the need for low-power circuits and power estimation is given along with different types of asynchronous circuit design styles. The chapter then details much of the work that has been done relating to this thesis and its contributions. Finally, the chapter concludes with an outline of the rest chapters of this thesis. 1.1 The Need for Low Power Due to shrinking technology sizes, the amount of power consumed by integrated circuits has risen steadily with each successive technology generation. While processors made decades ago required little in their cooling solutions, today’s processors with hundreds of millions of transistors require elaborate cooling solutions

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2 since they consume so much power. The trend for shrinking technology sizes and increasing power consumption can be seen in the international technology for semiconductors (ITRS) roadmap for the year 2006 in Table 1.1. Table 1.1 ITRS Road-Map, 2006 International Technology Road-Map for Semiconductors [56] Year of Production 2005 2006 2007 2008 2009 2010 2011 2012 2013 Gate Length (nm) 32 28 25 23 20 18 16 14 13 Transistors (mil.) 553 553 1106 1106 1106 2212 2212 2212 4424 Frequency (MHz) 5,204 6,7839,285 10,97212,36915,07917,658 20,065 22,980 Voltage (V) 0.9 0.9 0.8 0. 8 0.8 0.7 0.7 0.7 0.6 Battery (W) 2.8 3.0 3.0 3. 0 3.0 3.0 3.0 3.0 3.0 Max. Power (W) 167 180 189 198 198 198 198 198 198 One of the primary reasons for low power integrated circuit designs comes from the explosion in mobile devices. For example, from 1999-2001, the number of Americans adults with a cell phone increased by 29 percent [57]. Other mobile devices, like laptops, personal digital assistants, and MP3 players, have experienced similar growth in recent years. Since the performance of a mobile device is constrained by its battery lifetime, the need for low power integrated circuits within such devices is great. In devices that are not constrained by battery lifetime, the use of low power designs can lead to longer device lifetime and decreased power consumption. The operational lifetime of a device is directly related to the amount of power it consumes. As the amount of heat dissipated by a device decreases, the less the chance it can experience thermal-related failures, like electromigration. Electromigration can cause metal wires to expand and have electrical shorts. By decreasing the amount of power consumed by an integrated circuit, the amount of

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3 electricity consumed by the device is lowered. Since global warming is a major environmental issue and electricity generation leads to environmental and thermal pollution, by lowering the amount of electricity consumed in a device, the environment that all humans live in can be sustained. 1.2 Power Consumption The types of power consumption in an integrated circuit can be divided into two parts, static power consumption and dynamic power consumption. Static power consumption is dependent on the layout of transistors within a circuit as well as the process technology utilized, while dynamic power consumption is dependent on output transitions that lead to charging and discharging of load capacitances. The total power consumed by a circuit can be represented by the formula [41]: Ptotal = Pstatic + Pdynamic (1.1) 1.2.1 Static Power Consumption The first cause of static power consumption is subthreshold conduction through OFF transistors. While an ideal OFF transistor has no current flowing through it, the presence of subthreshold tunneling, leakage, and conduction leads to a small amount of current moving through the OFF transistor. The second cause of static power consumption is tunneling current through gate oxide. While silicon dioxide is a very good insulator, for technology processes that are smaller than 130nm, tunneling current becomes an important factor in static power consumption. The third cause of static power consumption is leakage through reverse-biased diodes. As the diffusion regions, wells, and substrate that make up the different parts of a transistor connect with each other, they create reverse-biased diodes. The static

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4 power consumption due to reverse-biased diodes is small compared to subthreshold conduction or gate tunneling and can generally be ignored. The fourth factor in static power consumption is contention current in ratioed circuits. For pseudo-nMOS circuit styles, there is a direct path from power to ground so contention current must be factored into the total static power consumption. The total static power consumption of a circuit is the product of the supply voltage and the total leakage current and is given by the formula [41]: Pstatic = IstaticVdd (1.2) 1.2.2 Dynamic Power Consumption The main cause of dynamic power consumption is the charging and discharging of gate load capacitances. For gates that are driving long interconnect wires, the parasitic capacitances present on these interconnect lines also contribute to the total load capacitance. When the output signal of a logic gate goes from low to high, the total load capacitance of a gate is charged with an energy equivalent to CV2 dd where C is the total load capacitance and Vdd is the supply voltage. When the output signal of a logic gate goes from high to low, the energy stored in the total load capacitance is discharged. The number of tim es an output signal transitions from zero to one is known as its switching activity, denoted by Given an operating frequency of f, the dynamic power consumption due to the switching activity of a circuit is [41]: f CV P2 dd dynamic (1.3) A secondary source of dynamic power consumption comes from short-circuit currents. Since an input signal cannot rise or fall between a logic one or zero

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5 instantaneously, the pMOS and nMOS transistors that are driven by the input signal will both be ON for a short period of time as the input signal transitions from Vdd to GND or vice-versa. Since both transistors ar e ON for a short period of time, this can lead to a short circuit between Vdd and GND. 1.3 Asynchronous Circuits Given the rise in transistor densities and power consumption by synchronous designs, asynchronous circuits are a low-power alternative. Since there is no clock to synchronize communication in an asynchronous circuit, the only parts of the circuit that will be contributing to dynamic power consumption are those that are doing a requested processing action. The followi ng subsections describe how communication in asynchronous circuits can be accomplished without a global clock as well as various implementation styles. For an exhaustive overview of asynchronous circuit design, the reader is directed to [54]. 1.3.1 Asynchronous Communication With the absence of a global clock to synchronize communication between logic elements, communication must be done another way. The different types of communication protocols that an asynchronous circuit can use include two-phase and four-phase handshaking [54]. Module A Module B request acknowledge data Figure 1.1 Asynchronous Communication An asynchronous communication example is shown in Figure 1.1. There is an acknowledge signal from Module B to Module A, and there are request and data

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6 signals from Module A to Module B. When Module A needs some processing action performed by Module B, it asserts the request signal. When Module B has finished performing this processing action, it asserts the acknowledge signal to show that the data is valid. Upon receipt of the asserted acknowledge signal, Module A lowers the request signal, and correspondingly, Module B lowers the acknowledge signal. This type of communication is known as two-phase handshaking and can be seen in Figure 1.2. Valid Data request acknowledge data Valid Data Figure 1.2 Two-Phase Handshaking While two-phase handshaking is simple to implement, it is not robust and can be prone to timing errors. An extension of two-phase handshaking, which is eventdriven, is four-phase handshaking, which is level-driven. During four-phase handshaking communication, data is valid the entire time the request line is asserted. The operation of four-phase handshaking can be seen in Figure 1.3. Valid Data request acknowledge data Valid Data Figure 1.3 Four-Phase Handshaking

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7 1.3.2 Asynchronous Circuit Implementation Styles Asynchronous circuit implementation styles can be categorized based on their timing model and how they encode data [54]. Huffman circuit styles assume that both wires and gates have delays. In Huffman circuits, also known as fundamental mode circuits, all gate and wire delays are limited by an upper bound. In Timed circuits, both wires and gates are assumed to have a delay. Unlike Huffman circuits, however, lower bounds in addition to upper bounds are placed on each of the delay elements in Timed circuits. Muller circuit styles place no bounds on gate delays and assume that wire delays are negligible. Quasi-delay insensitive (QDI) asynchronous circuits assume that aside from wires called isochronic forks, no delays are assumed on wires. Speed-independent and delay-insensitive implementation styles only assume that delays are present on gates; wires are assumed to have no delays. In asynchronous circuits, data can also be encoded using one signal line or two signal lines. Data that is encoded using one signal line is known as single-rail, while data that is encoded using two signal lines is known as dual-rail. In a dual-rail encoding, both signal lines must have the same logic value for data to be valid. Since both signal lines must have the same value for data to be valid, this makes dual-rail data encodings much more robust against timing errors and hence they are widely used in speed-independent and delay-insensitive asynchronous circuit design implementations. 1.4 Motivation As can clearly be seen from the previous sections, thermal constraints are an important factor in integrated circuit design. Asynchronous designs are inherently

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8 low power, so as synchronous circuit designs dissipate greater power with higher transistor densities, asynchronous circuits will become more attractive. However, asynchronous circuits are much more difficult to design than synchronous circuits. While there are many commercial CAD tools for synchronous designs, there are very few CAD tools fo r asynchronous circuit designs. Even though asynchronous circuits have been used in several large designs, like the AMULET embedded processor [58] and RAPPID instruc tion decoder [59], the lack of industrial tools prevents widespread adoption of the asynchronous design methodology. Within the integrated circuit design flow, timing, energy, power, and area constraints need to be checked at every stage. This work addresses the need for a universal tool or methodology that can accurately estimate the amount of energy consumed at the gate level in an asynchronous circuit. 1.5 Contributions This thesis makes several contributions. First, a new methodology for simulative energy estimation in the asynchronous domain has been developed. Second, as long as the asynchronous circuits used are based on the handshaking communication protocol, the proposed methodology can be used for any type of asynchronous circuit implementation style. As detailed in the related works, there are no simulative asynchronous energy estimation methodologies. Simulation of different asynchronous circuits is done using timed Petri nets. Petri nets can be used to model many things; in this case, they are used to model gate-level behavior of a circuit. With the introduction of time into a Petri net, gate and interconnect delays can be modeled; this allows the capture of

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9 energy dissipation due to glitches. CPN Tools is used to simulate the timed Petri net description of a circuit. This work is the first that integrates energy estimation into gate-level simulation of asynchronous circuits. This work uses the Balsa high-level asynchronous synthesis system to produce most of the circuits used. While Balsa can produce four-phase handshaking, quasi-delay insensitive, and dual-rail delay insensitive implementations of asynchronous circuits, this work uses four-phase handshaking circuits due to the extra gate overhead of delay-insensitive circuits and limitations present in the CPN Tools simulator. Any other tool besides Balsa that produces structural netlists of asynchronous circuits could also be integrated into this work. While very small asynchronous designs of less than 100 gates have been tested, the methodology proposed should be able to scale into the hundreds and thousands of gates; limitations currently present in the CPN Tools simulator prevent testing of larger asynchronous circuit designs. 1.6 Related Work This section describes the work related to this thesis. This includes power estimation in sequential and combinational circuits and energy estimation in asynchronous circuits. Power estimation methods exist at the architectural, RTL, transistor, and gate level. Furthermore, power estimation methods at the gate level can be subdivided into probabilistic and statistical methods.

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10 1.6.1 Sequential and Combinational Power Estimation Architectural Bagwell et al [6] Talarico et al. [7] Huang et al. [8] Givargis et al [9] Ludewig et al [10] Isci et al. [11] RTL Chang et al. [12] Sum et al. [13] Bruni et al. [14] Machado et al [15] Eiermann et al [16] Coburn et al. [17] Tran et al. [18] Chen et al. [19] Eiermann et al [20] Buyuksahin et al [44] Transistor Rossello et al [21] Shiue [22] Taherzadeh -S et al. [23] Jung et al. [24] Probabilistic Ramani et al. [25] Bhanja et al. [26] Cao [27] Dogandzic et al [28] Wright et al. [29] Bhanja et al [30] Freitas et al [35] Tsui et al. [42] Chou et al [43] Marculescu et al [45] Statistical Liu et al. [31] Evmorfopoulos et al [32] Liu et al. [33] Murugavel et al [34] Kozhaya et al [36] Garcia Ortiz et al [37] Sequential and Combinational Circuit Power Estimation Gate Figure 1.4 Taxonomy Diagram of Power Estimation at Different Levels of Abstraction in Sequential and Combinational Systems The latest methods in power estimation of embedded systems at the architectural level are given in references [6, 7, 8, 9, 10, 11]. By interfacing a Texas Instruments MSP430F149 microcontroller to a system’s output and supply voltage, Bagwell et al. [6] estimates the dynamic power consumption of a 340 megabyte IBM Microdrive inside the CompactFlash port of an IPAQ 3635. A power estimation method, proposed by Talarico et al. [7], takes into account a program’s instruction set, memory usage, and peripheral usage to analyze power in the different components of software, local bus, cache, system bus, main memory, peripheral bus, and peripherals. Given the prevalence of analog and digital devices in many systems, Huang et al. propose a method [8] to accurately estimate the power in many analogto-digital converters. To estimate the power in system-on-chip designs, a method proposed by Givargis et al. [9] evaluates power consumed in peripheral cores by combining simulation and gate-level power data. A technique proposed by Ludewig et al. [10] gathers interface signal transitions, and through the use of a statistical compression module, the captured transitions are stored in a compact data format; the

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11 data is then input into a rapid prototyping system, where the signals can be analyzed, and the amount of power consumed by the original system can be estimated. To estimate the power consumed in high-end processors, a technique proposed by Isci et al. [11] combine real power monitoring and component power estimation; the real power monitoring hardware consists of a clamp ammeter to measure current, a digital multimeter to read voltages on the clamp, a logger machine to collect the data, and a power monitor to display the captured data; the component power estimation part of this method takes the different components of a processor and their corresponding access rates to produce a power weight for each component. The latest methods and applications of RTL power estimation are given in references [12, 13, 14, 15, 16, 17, 18, 19, 20, 44]. To estimate the power dissipated in a security processor, Chang et al. [12] use the RTL level description of the processor to calculate the power dissipated by the processor during AES and RSA encryption and decryption. A method proposed by Sum et al. [13] uses an up-down encoding scheme with linear approximation to improve the estimation accuracy of RTL methods. Bruni et al. [14] propose a RTL power estimation flow that takes into account propagation delays; while conventional RTL power estimation flows assume a zero-delay model, the addition of delays allows the calculation of power consumption due to spurious transitions. Machado et al. [15] propose a technique to reduce power estimation complexity in VHDL-RTL designs by representing combinational logic blocks with Binary Decision Diagrams (BDDs). Eiermann et al. [16] propose several macromodeling techniques for RTL power estimation; the first modeling technique improves the accuracy of power dissipation caused by word-level

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12 switching activity by dividing all input bits into subwords, and the second modeling technique improves estimation of bit-level switching energy dissipation by adding an adjustment factor that is related to switching activity at the word-level. Coburn et al. [17] use specialized power estimation hardware to emulate the power characteristics of a target system. Tran et al. [18] estimate power consumption by weighting the total gate-count estimate against five elements of power consumption in a circuit: logic, on-chip memory, interconnection, clock distribution, and off chip driving I/O. Chen et al. [19] propose a technique to estimate RTL level power by predicting the node distribution, capacitance distribution, and entropy distribution of any Boolean function that is optimized for a minimal area implementation. Eiermann et al. [20] develop an efficient RTL power modeling t echnique for combinational logic blocks by only using word and bit level switching information. Buyuksahin et al. [44] propose a method for RTL power estimation that takes into account effects of interconnect loading between gates. The latest methods for estimating power at the transistor level are given in references [21, 22, 23, 24]. Rossello et al. [21] propose a model to accurately estimate the energy dissipated in domino CMOS gates by considering the internal capacitance switching and discharging currents of such circuits. Shiue [22] proposes a new analytical model for evaluating power at the transistor level; his analytical model consists of a -power law derived from physical MOSFET models, an analysis of future short-circuit power models, and the equation model from Berkeley BSIM3 manual. Taherzadeh-S et al. [23] develop a new model to calculate the power consumption of CMOS inverters; the model uses a modified MOSFET short channel

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13 n-th power law, and it calculates the short circuit current by using a linear interpolation scheme. Jung et al. [24] propose a new method for calculating shortcircuit power in static CMOS circuits by accurately deriving short-circuit current from the interpolation of peak points of act ual current curves, which are influenced by gate-to-drain coupling capacitance. The latest methods for probabilistic power estimation at the gate level are given in references [25, 26, 27, 28, 29, 30, 35, 42, 43, 45] Ramani et al. [25] use Bayesian networks to implement a probabilistic power estimation strategy that is nonsimulative, based on Importance Sampling, and scales efficiently with circuit complexity. Bhanja et al. [26] use Cascaded Bayesian Networks (CBNs) to estimate the switching activity of correlated input s within VLSI circuits; probabilistic consistency across the CBNs is maintained by a tree-dependent (TD) function, and the TD function is derived from a minimal weight spanning tree of switching activity that occurs along pairs of boundary signal lines. Cao [27] uses Bayesian inference and neural networks to develop a new tec hnique that enables an efficient table-lookup of power consumption of a circuit’s entire state and transition space. Dogandzic et al. [28] use sequential Bayesian networks and a Nakagami-m fading model to estimate the dynamic power of shadow channels in wireless communication networks. Wright et al. [29] propose a zero-delay probabilistic power estimation technique that uses a Signal Probability Computation procedure to calculate the signal probability for every node in a circuit; two representation models presented for a circuit include the Connective BDD and Integer Pair Representation. Bhanja et al. [30] use logicinduced directed acyclic graphs (LIDAGs) to model switching probability in a

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14 combinational circuit; LIDAGs can be mapped directly to a Bayesian network, which can be used to model the spatio-temporal correlation of switching activity on a given node. Freitas et al. [35] solve the Chapman-Kolmogorov equations for sequential state probabilities by using a discrete Markov chain input to accurately model all temporal and spatial correlations between internal nodes and primary inputs. A method by Tsui et al. [42] estimates glitch power by correlating the steady state value of two signal lines to the probability of them changing state; the greater the transition probability of an input signal, the greater the glitch power of the circuit. Chou et al. [43] propose a method to accurately capture switching activity in static and domino CMOS circuits by considering signal correlations and simultaneous switching among multiple input signal lines. Marculescu et al. [45] also present a technique to capture switching activity by considering spatiotemporal correlations under a zero-delay model. The latest methods for statistical power estimation at the gate level are given in references [31, 32, 33, 34, 36, 37]. Liu et al. [31] propose a new binary vector sequence generator based on a Markov chain (MC) model; the sequence generator accepts as input an average input probability, an average transition density, and a spatial correlation factor, and outputs a highly random and uniform vector sequence. Evmorfopoulos et al. [32] use a Monte Carlo approach based on extreme value theory to estimate the maximum power dissipation of CMOS VLSI circuits. Liu et al. [33] propose a new power macromodeling technique that accurately models input glitch propagation and its affects on energy dissipation within a circuit. Murugavel et al. [34] propose using sequential and recursive least squares estimation for average

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15 power estimation; least squares estimation converges faster than other statistical methods by attempting to minimize the error of the mean square value during each successive iteration of the algorithm. Kozhaya et al. [36] use blocks of randomly selected user-supplied input data to calculate upper and lower bounds on the power consumption of a circuit. Ortiz et al. [37] extend statistical power estimation methods to handle the nongaussian data models displayed by portable digital systems. 1.6.2 Asynchronous Energy and Power Estimation Asynchronous Energy and Power Estimation Probabilistic Gate-Level Lloyd et al [39] Beerel et al 48]Speed Independent [Burst Mode Beerel et al [40]This work Handshaking (All Types) Kudva et al [1] Self-TimedQuasi-Delay Insensitive Penzes et al [60] Salehi et al [61] Transistor-Level Simulative Behavioral-Level Quasi-Delay Insensitive Salehi et al [61] Niknahad et al [62] Figure 1.5 Taxonomy Diagram of Energy and Power Estimation Methods in Asynchronous Systems Kudva et al. [1] propose a method for estimating power in self-timed asynchronous circuits by conducting analysis on the discrete time Markov chain in the reachability graph obtained from the Petri net model of the circuit. In the absence of a global clock, a circuit is divided into three different parts: control elements (CBs), datapath elements (DBs), and data-dependent control blocks (PBs). Since the conventional average power estimation formula of: Pavg = * 2 12 dd loadVC f E(transitions) (1.4) where Vdd is the supply voltage, Cload is the output capacitance, f is the frequency of the circuit, and E(transitions) is the number of output transitions per global cycle, cannot be applied to a system with no global clock, the authors define switching

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16 energy per invocation. An invocation is a request and acknowledge transition pair between a PB and CB; self-timed circuits communicate using handshaking. The average switching energy consumed per invocation in a DB is given by: Einvocation = * 2 12 dd loadVCD(transitions) + Pdt (1.5) where Vdd is the supply voltage, Cload is the output capacitance, D(transitions) is the transition density, and Pdt is the energy consumed by the delay line in a PB. The average switching activity consumed per invocation in a PB is given by: Einvocation = * 2 12 dd loadVC D(transitions) + Pselect (1.6) where Pselect is the energy consumed when a request signal is followed by an acknowledgement signal. An example of the Petri net representation of a Muller C-element, a common asynchronous circuit element, is shown in Figure 1.6. The Petri net model representation is able to model interconnect delay through the Delay_wire annotations, and output transition delay is modeled through Delay_Celement annotation. The delays present in the wires and different circuit elements are used to build the timed reachability graph of the Petri net.

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17 ReqAck Celementout Delay_wireDelay_wire Delay_Celement Energy_Celement Figure 1.6 Petri Net Representation of Muller C-element [1] Lloyd et al. [39] propose a method for power estimation in speed-independent (SI) asynchronous circuits that uses invariant analysis of Petri net models using matrix representations. An SI circuit assumes all wire delays are negligible compared to gate output delays; thus, the proposed method is unable to model interconnect delay and glitch power caused by inputs arriving at different times. While this method avoids the state space explosion problem by using a matrix representation of a Petri net, the use of a gate delay model is only suited to SI circuits; the method described cannot be used to accurately estimate the power consumption in other types of asynchronous circuits. A method proposed by Beerel et al. [48] derives the sequential signal transition graph (SSTG) for a circuit. Markov chain analysis is done on the SSTG such that the average energy consumed per external signal transition is known.

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18 Beerel et al. [40] propose a method for energy estimation in burst-mode asynchronous circuits. Instead of operating on individual input signals, a burst-mode circuit operates with bursts of data; for example, an input burst will produce a corresponding output burst. A burst-mode circuit may have glitches within its internal signals, but its output is guaranteed to be hazard free. To characterize energy consumption in absence of a global clock, the average energy per output transition is considered, whose equation is given by [40]: * 2 1 ) (2 dd load gate gatesV T EnC (# of gate switches) (1.7) Furthermore, upper and lower bounds are placed on the number of gate switches so that the estimated energy dissipation falls within such bounds. There are three methods for estimating the energy consumed by quasi-delay insensitive circuits. Penzes et al. [60] develop a simulator called esim that operates on a transistor-level description of a QDI asynchronous circuit. They demonstrate their methodology by estimating the amount of energy consumed by an asynchronous MIPS R3000 microprocessor. Since many QDI asynchronous circuits are based on Pre-Charge Full/Half Buffer (PCFB/PCHB) logic, Salehi et al. [61] develop a methodology to estimate energy consumption by measuring the transition count of behavioral-level and transistor-level descri ptions of QDI circuits using verilog HDL models. A simulative method developed by Niknahad et al. [62] estimates the power consumption in QDI circuits at the behavioral-level by counting the number of read and writes in a circuit.

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19 1.7 Thesis Overview Chapter 2 provides a background on Petri nets and previous related work that uses Petri nets for power estimation. Chapter 3 provides details of the Petri net developed to accurately capture the behavior of different gates. Chapter 4 provides an overview of the power estimation framework used. Chapter 5 provides experimental results, and Chapter 6 is devoted to conclusions.

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20 CHAPTER 2 PETRI NET FRAMEWORK The first part of this chapter gives a formal introduction to Petri nets and their operation. The second part of this chapter describes the previous related work that has been done using Petri nets for power estimation. 2.1 Petri Net Basics A Petri net model [38] is composed of a set of places and a set of transitions. The transitions represent an action the model can take, while places represent the state of a model. To connect the places and transitions in a Petri net, a set of directed arcs are used. There are two types of directed arcs in a Petri net, input arcs and output arcs. Input arcs are directed arcs that connect places to transitions. Output arcs are directed arcs that connect transitions to places. A transition is enabled to fire if all places that are connected to the input arcs of the transition have a token. Once a transition fires, tokens from input places are consumed and deposited on places connected by the output arcs from the transition. The firing of a transition represents a change of state for the Petri net. The firing of transitions in a Petri net is non-deterministic, which means if multiple transitions are enabled, then they can fire in any random order. The initial marking of a Petri net denotes which places of the Petri net are initialized with tokens. A Petri net can be formally defined: PN = {P, T, A, P0} (2.1)

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21 where: P = set of places T = set of transitions A = set of arcs from places to transitions or transitions to places P0 = initial set of marked places There are many different types of Petri nets: colored, timed, hierarchical, deterministic, predicate/transition, and stochastic. The basis of this work comes from colored Petri nets. Colored Petri nets (CP-Nets) are Petri nets that have been expanded to include color sets, arc expressions, and guard functions. The color sets in a CP-Net control the type of tokens a place can store as well as determining the operation and functionality of the overall Petri net. Arc expressions attached to arcs in a CP-Net can be used to control the binding of token values when a transition fires. The guard functions in a CP-Net are Boolean expressions attached to transitions. A guard expression must evaluate to true for a transition in a CP-Net to be enabled to fire. A CP-Net also has a node function to map arcs to source and destination nodes. For input arcs in a CP-Net, source nodes are places, and destination nodes are transitions. For output arcs in a CP-Net, source nodes are transitions, and destination nodes are places. The initialization function of a CP-Net determines which places have tokens when the net is initialized. A CP-Net can be formally defined [38]: CPN = (, P, T, A, N, C, G, E, I) (2.2) where: is a finite set of color sets P is a finite set of places

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22 T is a finite set of transitions A is a finite set of directed arcs such that A T A P T P N is a node function that is defined from A into P. T T P C is a color function that is defined from P into G is a guard function E is an arc expression function I is an initialization function The following definitions [55] are necessary to define the behavior of a CP-Net: Definition 1 [55]: A multi-set m, over a non-empty set S, is a function ] [ N S m which is represented as a formal sum: S ss s m)' ( (2.3) where SMS denotes the set of all multi-sets over S, N represents the set of all nonnegative integers, the non-negative integers } | ) ( { S s s m are the coefficients of the multi-set, and m s iff 0 ) ( s m Definition 2 [55]: For all tT and for all pairs of nodes TxP) T (P ) x (x2 1 : P} {t} {t} P N(a) | A {a A(t) (2.4) Var(E(a))} v : A(t) a Var(G(t)) v | {v Var(t) (2.5) )} x (x N(a) | A {a ) x A(x2 1 2 1 (2.6) ) x A(x a 2 12 1E(a) ) x E(x (2.7) Definition 3 [55]: A binding of a transition t is a function b defined on Var(t) such that:

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23 Type(v) b(v) : Var(t) v (2.8) G(t) (2.9) where B(t) is the set of all bindings for t, G(t) denotes the evaluation of the guard expression G(t) in the binding b, and Type(v) denotes the type of a variable v Definition 4 [55]: A token element is a pair (p, c) where p P and cC(p), while a binding element is a pair (t, b) where t T and b B(t). The set of all token elements is denoted by TE while the set of all binding elements is denoted by BE. A marking is a multi-set over TE while a step is a non-empty and finite multi-set over BE. The initial marking M0 is the marking which is obtained by evaluating the initialization expression: I(p))(c) ( c) p, ( M : TE c) (p,0 (2.10) The initialization expression that determines the initial marking M0 for all places p and all colors c is given by (I(p))(c). The sets of all markings and steps are denoted by M and Y, respectively. Definition 5 [55]: A step Y is enabled in a marking M iff the following property is satisfied: Y b) (t,M(p) b t) E(p, : P p (2.11) where b t) E(p, denotes an expression evaluation that removes tokens from place p and fires transition t with the binding b. Once property 2.7 is satisfied, (t,b) and t are said to be enabled. When a step Y is enabled in a marking M1 it may occur, changing the marking M1 to another marking M2 defined by:

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24 Y b) (t, Y b) (t, 1 2b t) E(p, ) b t) E(p, (p) M ( p) ( M : P p (2.12) M2 is directly reachable from M1. This is written 2 1M Y M Definition 6 [55]: A finite occurrence sequence is a sequence of markings and steps: 1 n n n 3 2 2 1 1M Y M . M Y M Y M (2.13) such that nN, and 1 i i iM Y Mfor all i {1, 2, …, n} M1 is the start marking, Mn+1 is the end marking and n is the length. Analogously, an infinite occurrence sequence is a sequence of markings and steps: . M Y M Y M3 2 2 1 1 (2.14) such that 1 i i iM Y M for all 1. i A marking M"is reachable from a marking M' iff there exists a finite occurrence sequence starting in M' and ending in M". The set of markings that is reachable from M' is denoted by M'. A marking is reachable iff it belongs to M0. CP-Nets can also be augmented to include hierarchy and time. Hierarchical CP-Nets (HCPNs) are the foundation for a new cl ass of Petri net that can be used to model an asynchronous circuit. To accurately describe the behavior of HCPNs, the following definitions are needed: Definition 7 [38]: A time set (TS) is defined as the set of all non-negative real numbers, TS = xR, where x represents the time instant and R is the set of all real numbers. Definition 8 [38]: For each transition t T, Var(t) is the set of variables associated with transition t given as,

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25 Var(E(a))} v : A(t) a Var(G(t)) v | {v Var(t) : T t (2.15) where, v is the variables in the Petri net, Var(G(t)) is the set of variables associated with the Guard-expression (G), a is the arcs in the Petri net, A(t) is the set of arcs associated with transition t, and E(a) is the arc-expression associated with arc a. 2.2 Power Estimation Using Petri Nets Since this work uses Petri nets to model asynchronous systems at the gate level, it draws heavily on the work done previously in [3, 4] for power estimation in combinational and sequential circuits. This section describes the work done in [3, 4, 5]. Murugavel et al. [3] develop the foundation for power estimation in combinational circuits through the use of Hierarchical Colored Hardware Petri nets (HCHPNs). A basic HCHPN is a structural model of a gate. HCHPNs are derived from colored Petri nets (CPNs). A HCHPN [3] is defined as the set HCHPN = (PG, ST, GT, TI, TO, TC, RP) (2.16) where: PG is a finite set of non-hierarchical CPN T ST where ST is a set of substitution transitions (super nodes) (T GT and ST GT = ), where GT is the set of gate transitions TI is the time set defined as the set of all nonnegative real numbers, TI = x R, where x represents the time instant and R the set of all real numbers TO is the set of all possible colored tokens TC is the code segment function. It is defined from T into the set of functions P RP is a set of restricted places, which can hold only a finite numbers of tokens in them.

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26 PI1 PI2 P 0 P1 P2 P3 P4 P5 POw w w w w w w w w w w w w w w w : place: transition : restricted place : gate transition W : (# of tokens, type of token = 0 or 1) w0: (# of tokens, type of token = 0 only) w1: (# of tokens, type of token = 0 only) T0 T1 T2 T3 T4 T5 T6 T7w0 w0 w0 w0 w0 w0 w0 w0 w1 w1 w1 w1 w1 w1 Figure 2.1 HCHPN Structure of NAND Gate [3] As an example of how HCHPNs can be used to accurately represent a gate, including gate delays and interconnect delays, the reader is referred to Figure 2.1, which shows the HCHPN representation of a NAND gate. Places {PI1, PI2} correspond to the two input nodes of a NAND gate, and place PO corresponds to the output node of a NAND gate. Input glitches are captured using places {P0, P1, P2} and transitions {T0, T1}. Since places {P0, P1, P2} are restricted places, the arrival of a new token will always overwrite an existing token. The gate function is incorporated into the transition {T2}; a Boolean expression that corresponds to the function of the gate must evaluate to true before transition T2 can be fired. The switching activity of the gate is captured by using places {P3, P4, P5} and transitions {T3, T4, T5, T6}. To determine whether the gate output is rising, falling, or remaining the same, place {P3} stores the current state of the gate, and place {P5}

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27 stores the previous state of the gate. Transitions {T3, T4, T5, T6} correspond to different types of switching activity within a gate: T3 fires if the output of the gate remains at logic ‘0’ T4 fires if the output falls from logi c ‘1’ to logic ‘0’ (power dissipation) T5 fires if the output rises from logic ‘0’ to logic ‘1’ (power consumption) T6 fires if the output of the gate remains at logic ‘1’ The output of the transitions {T3, T4, T5, T6} is stored in places {P4, P5}. The temporal correlation between different tokens is modeled using transition {T7} and places {P4, P5}. The structural HCHPN model of the NAND gate serves as the basis for future Petri net development work; it is able to accurately model the delay, gate function, switching activity, glitch power, and overall power consumption present within the gate. Murugavel et al. [4] extend the HCHPN described previously to handle sequential circuits. Since sequential circuits are basically combinational circuits with delayed output feedback and flipflops, it is not hard to modify a combinational HCHPN to account for the unique temporal and spatial relationships present within a sequential circuit. The tool flow for power estimati on using HCHPNs is shown in Figure 2.2. Since the HCHPN [3] serves as the foundation for future Petri net modeling of asynchronous systems, it is logical that the tool flow for switching activity and energy estimation would be similar to the ones used in references [3, 5].

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28 Gate-level netlist input GSG HCHPN HCHPN simulation Switched capacitance Power calculator Delay Calculator R-C input layout Gate delays Input patterns Figure 2.2 Tool Flow for HCHPN Power Estimation

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29 CHAPTER 3 PETRI NET MODELING OF ASYNCHRONOUS CIRCUITS FOR ENERGY ESTIMATION In this chapter, a new type of Petri net for energy estimation in asynchronous circuits is presented. The first part of this chapter gives the formal definition for the new type of Petri net. The second part of this chapter describes how basic logic gates can be modeled using the new type of Petri net. Within each subsection of the second part of this chapter, detailed firing examples are given to show how the Petri net operates. The third and final part of this chapter describes how an asynchronous circuit can be transformed into the new type of Petri net while preserving its behavior and timing information. 3.1 Hierarchical Colored Asynchronous Hardware Petri Nets To develop a new type of Petri net that can accurately model the gate delays and interconnect delays present in a circuit, the factor of time must be added. Also, since communication in asynchronous circuits is predominantly based on request/acknowledge handshaking, the number of invocations in a circuit must also be modeled. To accomplish these goals, a new type of Petri net, called Hierarchical Colored Asynchronous Hardware Petri net (HCAHPN), has been developed. The features of HCAHPNs are:

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30 Gate delays and interconnect delays of a circuit are accurately captured Special substitution transitions count the number of request and acknowledge signal pairs to determine the number of invocations in a circuit When gate transitions fire in a HCAHPN, it is the equivalent of a gate evaluating to a logic one or zero Each token in a HCAHPN has a time stamp. The token is only available to be consumed by transitions if its time value is equal to or less than the current simulator time. To accurately model each gate in a high-level circuit, a HCAHPN has a set of substitution transitions. Each substitution transition in a HCAHPN corresponds to a subnet that accurately models a primitive gate. A signal propagating through a logic gate in a circuit is analogous to a token propagating through a substitution transition in a HCAHPN. In addition to substitution transitions, the HCAHPN also has a set of gate transitions that take into account the value of input tokens and produce an appropriate output token. The logic behavior of a gate is accurately modeled by gate transitions within a HCAHPN. Depending on wh at type of signal a gate is driving, a substitution transition can be classified as a request or acknowledge substitution transition. In these special types of substitution transitions, the switching activity recorded within the subnet also serves to record the number of request and acknowledge signal changes. A token in a HCAHPN is defined by the tuple (p, v, n, t), where p is the place of the token, v is the color of the token, n is the port number of the token, and t is the time stamp value of the token. The color value of the token is used to represent the logic value of a signal. The port number of a token is used to

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31 determine from which input signal to a gate the token came from; it can also be used to denote a gate output signal or control the enabling of transitions. Code segments can also be attached to arcs and transitions to control the flow and values of tokens. A HCAHPN can be defined mathematically as follows. Definition 9: A HCAHPN can be defined as a tuple: HCAHPN = (CP, ST, GT, TS, CT, CS, RST, AST) (3.1) where: CP is a set of non-hierarchical CP-Nets ST is a set of substitution transitions such that T ST GT is a set of gate transitions such that T GT and ST GT = TS is the time set as defined previously in definition 7 of section 2.1 CT is the set of all possible colored tokens denoted by the tuple (p, v, n, t), where pP, v is the value of the token, n is the port number of the token, and tTS CS is the code segment function. It maps code segments into the set of gate transitions GT, the set of guard functions G, and the set of arc expressions E. RST is the set of request substitution transitions such that ST RST and AST RST = AST is the set of acknowledge substitution transitions such that ST AST and AST RST = The following definitions are necessary to define the various features of a HCAHPN, which is useful for modeling asynchronous circuits: Definition 10: For each CP-Net cp CP, the following characteristics hold:

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32 cp(a)) (CP(a) cp(a) (3.2) cp(p)) (CP(p) cp(p) (3.3) cp(t)) (CP(t) cp(t) (3.4) where cp(a) denotes the set of arcs associat ed with cp, CP(a) denotes the set of all arcs in CP, cp(p) denotes the set of places a ssociated with cp, CP(p) denotes the set of all places in CP, cp(t) denotes the set of transitions associated with cp, and CP(t) denotes the set of all transitions in CP. Definition 11: A token element in a HCAHPN is a tuple (p, v, n, t) where p P, vC(p), nC(p), and tTS. Definition 12: A Firing (t, x, y) specifies the possibility of the firing of transition t by removing tokens from the input places x attached to transition t and depositing tokens to the output places y attached to transition t. The set of all firings is the Firing set FS. Definition 13: The enabling time F of a firing (t, x, y) is the maximal value of the time-stamps of all the tokens in the set of input places x when transition t fires. xCT t) n, v, (p,max b) a, F(t, (3.5) Definition 14: If a transition t is enabled and the global clock is greater than or equal to the enabling time, it can fire. If more than one transition is enabled to fire at the same time, one transition is non-deterministically selected from the set of enabled transitions to fire. The global clock does not increment until all enabled transitions at the given time have fired.

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33 3.2 HCAHPN Low-Level Modeling A B Z Figure 3.1 AND Gate As an example of how HCAHPNs can be used to model different types of logic gates, the structure of a HCAHPN for the AND gate in Figure 3.1 is shown in Figure 3.2. TG IP1 IP2 IT2 IT1 P0 PS1 PS2 P1 DC0 DC1 P2 P3 OTP4P5 TF0 TF1 OP place transition TG gate transition Gate FunctionalitySwitching ActivityToken Filter Figure 3.2 HCAHPN structure of AND Gate A HCAHPN structure can be divided into three parts: gate functionality, switching activity, and token filter. A detailed explanation of each of these different parts of the HCAHPN is provided later in this section. While the structure of a two input AND gate has been presented, th e modeling of OR, NAND, NOR, BUF, INV, and DFF as a HCAHPN is very similar.

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34 3.2.1 Gate Functionality Substructure IP1 IP2 P0 IT2 IT1 (v, p) (v, p) TG PS1 PS2 P1 input (v, p, v1, v2) output (vO, pO); action let in if p = 1 andalso (v = 1 andalso v2 = 1) then (1, 0) else if p = 2 andalso (v =1 andalso v1 = 1) then (1, 0) else (0, 0) end; (v, 1) (v, 2) (v, p) if p = p1 then (v, 1) else (v1, 1) if p = p2 then (v, 2) else (v2, 2) (v1, p1) (v2, p2) (vO, pO) @+ 159 (v3, p3) @+ 158 Gate Transition Code Segment place transition TG gate transition v, v1, v2, v3, vO: To ken Logic Variables p, p1, p2, p3, pO: Token Port Number Variables @+ arc delay Figure 3.3 Gate Functionality Structure Detail of AND Gate The details of the gate functionality Petri net substructure of a HCAHPN model for an AND gate are shown in Figure 3.3. The variables present on the different arcs are used to bind token values. It is important to note that the scope of each variable is local to each transition. For example, the scope of the variable set (v, p) attached to transition {IT2} is different from the variable set (v, p) attached to transition {TG}. Once all variables attached to a transition can be bound to a value from their respective tokens, that transition is enabled to fire. The places {IP1, IP2} are used to model the two input ports of an AND gate. The transitions {IT1, IT2} are used to propagate tokens from places {IP1, IP2} to place {P0}. The places {PS1, PS2} are used to store the current value of inputs A and B respectively. Place {P0} is used to store the value of current inputs to the

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35 circuit. By the structure of the Petri net, the capacity of places {PS1, PS2} is limited to one token. Through the use of a code segment function, transition {TG} is able to use the value of tokens from places {P0, PS1, PS2} to model the functionality of the gate. The delay on the arc from transition {TG} to place {P1} is used to model the intrinsic gate delay of the AND gate. While place {P0} can store multiple tokens, a token is only available once its time stamp is equal to the simulator time. If there are multiple input tokens on place {P0} with successively increasing time stamp values, the simulator will evaluate each token deterministically as simulator time increases and each token becomes available to the simulator. If two tokens have a time stamp value of i at place {P0}, then at time i the transition {TG} will non-deterministically choose one token to consume. At time i + 1, the transition {TG} will fire again to consume the second token at place {P0}. Since intrinsic gate delay is much greater than 1 time unit and the token at place {P1} is not available until the gate output finishes transitioning, this very small non-deterministic behavior is acceptable. Since there must be tokens present for a transition to fire, the initialization function is used to initialize tokens at places {PS1, PS2, P1}. Since the tuple (v, p, n, t) is the set of values a token can take, the following is a formal description of how the initialization function initializes tokens at places {PS1, PS2, P1} PS1 = (0, 1, PS1, 0) PS2 = (0, 2, PS2, 0) P1 = (0, 100, P1, 0)

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36 The port number value of a token serves two purposes. First, it is used to represent from which input port a token came from. For example, as a token propagates from place {IP1} to place {P0} through transition {IT1}, its port value is forced to 1. Second, its value is used as part of a guard function on transitions. A port number value of 0 designates that the token is an output. Place P1 is initialized with a token whose port number value is 100. Since there are no gates with one hundred input ports, this arbitrary value was chosen to control the firing of transitions in the switching activity substructure of the HCAHPN. Within CPN Tools, delays can be specified on transitions, input arcs, and output arcs. While an HCHPN uses transition delays to model intrinsic gate delays, a HCAHPN uses delays on output arcs to model intrinsic gate delays. A delay on an arc or transition updates the time value of all output tokens attached to the arc or transition. A token is only available to be used by a transition if its time value is less than or equal to the current simulator time. If gate delays were specified on transition {TG}, then the firing of the Petri net could become non-deterministic. For example, suppose transition {TG} has just fired and the time value of tokens at places {P0, PS1, PS2, P1} is 159 time units greater than the current simulator time. The tokens at these places are not available to be used by any transitions until the current simulator time has incremented an additional 159 time units. During the incrementing of simulator time, places {IP1, IP2} each receive a token. Once 159 time units have passed, the tokens at places {P0, PS1, PS2, P1} are available to be used by transitions. This situation leads to a non-deterministic firing of the transitions {IT1,

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37 IT2}. Since it is very important to preserve the temporal correlation of inputs in the HCAHPN, such a situation is unacceptable. The HCAHPN is structured such that only complete transitions of a gate are modeled. The delays present on both arcs that link transition {TG} and {P1} are used to accomplish this. A delay present on an output arc updates the time value of a token by adding the delay value present on the arc to the current simulator time. A delay present on an input arc to a transition specifies that a transition can consume tokens from a place before a token’s time value in that place equals the current simulator time. Since the intrinsic gate delay of a two input AND gate is 159 time units, this is captured in the output arc delay from transition {TG} to place {P1}. Note however, the input arc delay of 158 time units on the ar c from place {P1} to transition {TG}. If a new input arrives to the AND gate as the simulator increments 159 time units, the output token on place {P1} will be consumed and a new output token will be produced that reflects the current state of the output. For all gates modeled, if an intrinsic gate delay is modeled as n time units on an output arc, then the delay present on an input arc will be n – 1 time units. While this work uses a real delay model to capture glitch power, the difference in arrival time between two different inputs must be greater than the intrinsic gate delay of a gate in order for such a glitch to be captured. The code segment function attached to transition {TG} is used to model the functionality of the AND gate. The values of tokens from places {PS1, PS2} are evaluated by the code segment function and the appropriate value for the output token

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38 is chosen. The coding language used to describe the code segment function is Standard ML. 3.2.1.1 Gate Functionality Firing Example This section provides an example of how the gate functionality substructure of the AND gate HCAHPN evaluates tokens. (v, p) (v, p) TG (v, 1) (v, 2) (v, p) if p = p1 then (v, 1) else (v1, 1) if p = p2 then (v, 2) else (v2, 2) (v1, p1) (v2, p2) (vO, pO) @+ 159 (v3, p3) @+ 158 0 100 0 2 0 1 place transition TG gate transition v, v1, v2, v3, vO: Token Logic Variables p, p1, p2, p3, pO: Token Port Number Variables @+ arc delay IT2 IT1 IP1 IP2 P0 P1 PS2 PS1 Figure 3.4 Gate Functionality Step 0 In Figure 3.4, the gate functionality substructure has just been initialized. The current simulator time is 0. Formally, the values of tokens at all places are: PS1 = (0, 1, PS1, 0) PS2 = (0, 2, PS2, 0) P1 = (0, 100, P1, 0)

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39 (v, p) (v, p) TG (v, 1) (v, 2) (v, p) if p = p1 then (v, 1) else (v1, 1) if p = p2 then (v, 2) else (v2, 2) (v1, p1) (v2, p2) (vO, pO) @+ 159 (v3, p3) @+ 158 0 100 0 2 0 1 1 0 IP2 place transition TG gate transition v, v1, v2, v3, vO: Token Logic Variables p, p1, p2, p3, pO: Token Port Number Variables @+ arc delay IP1 P0P1 IT2 IT1 PS2 PS1 Figure 3.5 Gate Functionality Step 1 At simulator time 160, a new input token arrives at place IP1, as shown in Figure 3.5. With the arrival of token at place {IP1}, transition {IT1} is enabled to fire. On the inbound arc to the transition, variable v is bound to 1, and variable p is bound to 0. On the outbound arc from the transition, the logic value of the token remains the same, but the port value is changed from 0 to 1. The simulator time is still 160. Formally, the value of tokens at all places are: PS1 = (0, 1, PS1, 0) PS2 = (0, 2, PS2, 0) P1 = (0, 100, P1, 0) IP1 = (1, 0, IP1, 160)

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40 (v, p) (v, p) TG (v, 1) (v, 2) (v, p) if p = p1 then (v, 1) else (v1, 1) if p = p2 then (v, 2) else (v2, 2) (v1, p1) (v2, p2) (vO, pO) @+ 159 (v3, p3) @+ 158 0 100 0 2 0 1 1 1 place transition TG gate transition v, v1, v2, v3, vO: Token Logic Variables p, p1, p2, p3, pO: Token Port Number Variables @+ arc delay IP2 IP1 IT2 IT1 PS1 PS2 P0P1 Figure 3.6 Gate Functionality Step 2 Figure 3.6 shows the status of the Petri net after transition {IT1} has been fired. After propagating to place {P0}, gate transition {TG} is enabled to fire. Since the port value of the token at place {P0} is 1, the code expression on the arc from transition {TG} to place {IP1} will overwrite the previous token stored at place {IP1} with the new token from place {P0}. Since only one of the inputs has a value of 1, the output of the gate remains at 0. The simulator time is still 160. Formally, the values of tokens at all places are: PS1 = (0, 1, PS1, 0) PS2 = (0, 2, PS2, 0) P1 = (0, 100, P1, 0) P0 = (1, 1, P0, 160)

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41 (v, p) (v, p) TG (v, 1) (v, 2) (v, p) if p = p1 then (v, 1) else (v1, 1) if p = p2 then (v, 2) else (v2, 2) (v1, p1) (v2, p2) (vO, pO) @+ 159 (v3, p3) @+ 158 0 0 0 2 1 1 place transition TG gate transition v, v1, v2, v3, vO: Token Logic Variables p, p1, p2, p3, pO: Token Port Number Variables @+ arc delay IP2 IP1 IT1 IT2 PS2 PS1 P0P1 Figure 3.7 Gate Functionality Step 3 The status of the Petri net after gate transition {TG} fires is shown in Figure 3.7. Note that the time stamp of token at place {P1} has been incremented by 159 time units. It will be available to the simulator at time 319. The input arc from place {P1} to transition {TG}, however, allows transition {TG} to access the token at place {P1} up to 158 time units before the simulator time equals 319. The simulator time is still 160. Formally, the values of tokens at all places are: PS1 = (1, 1, PS1, 160) PS2 = (0, 2, PS2, 160) P1 = (0, 0, P1, 319)

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42 (v, p) (v, p) TG (v, 1) (v, 2) (v, p) if p = p1 then (v, 1) else (v1, 1) if p = p2 then (v, 2) else (v2, 2) (v1, p1) (v2, p2) (vO, pO) @+ 159 (v3, p3) @+ 158 0 0 0 2 1 1 1 0 place transition TG gate transition v, v1, v2, v3, vO: Token Logic Variables p, p1, p2, p3, pO: Token Port Number Variables @+ arc delay IT2 IT1 IP2 IP1 PS2 PS1 P1 P0 Figure 3.8 Gate Functionality Step 4 When simulator time reaches 300 however, a new input token arrives at input place {IP2} as shown in Figure 3.8. Transition {IT2} is enabled to fire. On the inbound arc to the transition, variable v is bound to 1, and variable p is bound to 0. On the outbound arc from the transition, the logic value of the token remains the same, but the port value is changed from 0 to 2. The simulator time is 300. Formally, the values of tokens at all places are: PS1 = (1, 1, PS1, 160) PS2 = (0, 2, PS2, 160) P1 = (0, 0, P1, 319) IP2 = (1, 0, IP2, 300)

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43 (v, p) (v, p) TG (v, 1) (v, 2) (v, p) if p = p1 then (v, 1) else (v1, 1) if p = p2 then (v, 2) else (v2, 2) (v1, p1) (v2, p2) (vO, pO) @+ 159 (v3, p3) @+ 158 0 0 0 2 1 1 1 2 place transition TG gate transition v, v1, v2, v3, vO: Token Logic Variables p, p1, p2, p3, pO: Token Port Number Variables @+ arc delay PS2 PS1 P1 P0 IT2 IT1 IP2 IP1 Figure 3.9 Gate Functionality Step 5 After transition {IT2} fires, the gate transition {TG} is enabled to fire, as shown in Figure 3.9. Since the port value of the token at place {P0} is 2, the code expression on the arc from transition {TG} to place {IP2} will overwrite the previous token stored at place {IP2} with the new token from place {P0}. Since both of the inputs have a value of 1, the output of the ga te will transition to 1. The simulator time is still 300. The delay expression on the input arc from place {P1} to transition {TG} allows transition {TG} to consume the token at place {P1}. Formally, the values of tokens at all places are: PS1 = (1, 1, PS1, 160) PS2 = (0, 2, PS2, 160) P1 = (0, 0, P1, 319) P0 = (1, 2, P0, 300)

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44 (v, p) (v, p) TG (v, 1) (v, 2) (v, p) if p = p1 then (v, 1) else (v1, 1) if p = p2 then (v, 2) else (v2, 2) (v1, p1) (v2, p2) (vO, pO) @+ 159 (v3, p3) @+ 158 1 1 1 2 1 0 place transition TG gate transition v, v1, v2, v3, vO: Token Logic Variables p, p1, p2, p3, pO: Token Port Number Variables @+ arc delay PS2 PS1 P1 P0 IT1 IT2 IP2 IP1 Figure 3.10 Gate Functionality Step 6 The status of the Petri net after gate transition {TG} fires is shown in Figure 3.10. The gate output begins to rise as the logic values of both inputs are now 1. At simulator time value 459, the token at place {P1} can propagate through the switching activity substructure of the HCAHPN as its time stamp value will equal simulator time. As simulator time increments to 459, no new input tokens arrive at input places {IP1, IP2}. Formally, the values of tokens at all places are: PS1 = (1, 1, PS1, 300) PS2 = (1, 2, PS2, 300) P1 = (1, 0, P1, 459)

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45 3.2.2 Switching Activity Substructure (v1, p1) (v1, p1) (v, p) (v, p) (v, p) (v, p) (v, 100) (v, 100) (v, p) (v, p) (v1, p1) (v, p) place transition v, v1: Token Logic Variables p, p1: Token Port Number Variables DC0 DC1P3 OT P2 P4 P1 Figure 3.11 Switching Activity Structure Detail of AND Gate The details of the switching activity Petri net substructure of a HCAHPN model for an AND gate are shown in Figure 3.11. The current output of the gate is stored at place P1. Once the time value of the token at place P1 is equal to the current simulator time, it is available to transitions {DC0, DC1}. Note the arcs from transitions {DC0, DC1} to place {P1} that set the port value of a token to 100. Part of the guard functions present on the transitions {DC0, DC1} specifies that the transitions are only enabled if the port value of a token present at place {P1} is not equal to 100. Also, a token must be present on place {P4} for transition {TG} to be enabled.

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46 The switching activity substructure of a HCAHPN is similar to that of an HCHPN. Place {P3} is used to hold the previous output of the gate. Once an appropriate transition from the set {DC0, DC1} has fired, the current state of the gate will be reflected at the token on place {P2}. After transition {OT} fires, the token value present at place {P2} is propagated to places {P3, P4}. The previous state of the gate that is reflected by the value of token at place {P3} before the firing of transition {OT} is overwritten by the value of the token from place {P2} after transition {OT} fires. Each transition from the set {DC0, DC1} is enabled depending on the current state and previous state of the output of the gate. Transition DC0 fires if the gate output remains the same Transition DC1 fires if the gate output is rising or falling Within CPN Tools, monitors are attached to transition DC1 to record the switching activity of the output of the gate. For gates that are driving request or acknowledge signals, the monitor attached to transition DC1 also serves to record the number of request and acknowledge signal assertions. There is no token present at place P2 when the Petri net is initialized. The initialization function initializes tokens at places {P1, P3} to the following values: P1 = (0, 100, P1, 0) P3 = (0, 0, P3, 0)

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47 3.2.2.1 Switching Activity Firing Example This section provides an example of how the switching activity substructure of the AND gate HCAHPN evaluates tokens. (v1, p1) (v1, p1) (v, p) (v, p) (v, p) (v, p) (v, 100) (v, 100) (v, p) (v, p) (v1, p1) (v, p) place transition v, v1: Token Logic Variables p, p1: Token Port Number Variables DC0 DC1P3 OT P2 P4 P1 0 100 0 0 Figure 3.12 Switching Activity Step 0 In Figure 3.12, the switching activity substructure has just been initialized. The current simulator time is 0. Formally, the values of tokens at all places are: P1 = (0, 100, P1, 0) P3 = (0, 0, P3, 0)

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48 (v1, p1) (v1, p1) (v, p) (v, p) (v, p) (v, p) (v, 100) (v, 100) (v, p) (v, p) (v1, p1) (v, p) place transition v, v1: Token Logic Variables p, p1: Token Port Number Variables DC0 DC1P3 OT P2 P4 P1 1 0 0 0 Figure 3.13 Switching Activity Step 1 At simulator time 459, a new token is available to the simulator at place {P1}, as shown in Figure 3.13. Since the new t oken’s output value is different from the previous output token value stored at place {P 3}, transition {DC1} is enabled to fire. The simulator time is 459. Formally, the values of tokens at all places are: P1 = (1, 0, P1, 459) P3 = (0, 0, P3, 0)

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49 (v1, p1) (v1, p1) (v, p) (v, p) (v, p) (v, p) (v, 100) (v, 100) (v, p) (v, p) (v1, p1) (v, p) place transition v, v1: Token Logic Variables p, p1: Token Port Number Variables DC0 DC1P3 OT P2 P4 P1 1 100 0 0 1 0 Figure 3.14 Switching Activity Step 2 After transition {DC1} fires, as shown in Figure 3.14, the current output token of the gate is sent to place {P2}. Since there must be a token at place {P1} for transition {TG} to fire, a new token is also deposited at place {P1}. The port value of the new token at place {P1} is set to 100 to prevent transitions {DC0, DC1} from being enabled. The monitor attached to transition {DC1} is also incremented to record the switching activity of the gate. The simulator time is 459. Formally, the values of tokens at all places are: P1 = (1, 100, P1, 459) P2 = (1, 0, P2, 459) P3 = (0, 0, P3, 459)

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50 (v1, p1) (v1, p1) (v, p) (v, p) (v, p) (v, p) (v, 100) (v, 100) (v, p) (v, p) (v1, p1) (v, p) 1 0 1 100 1 0 place transition v, v1: Token Logic Variables p, p1: Token Port Number Variables P3 DC1 P1 DC0 P2 OT Figure 3.15 Switching Activity Step 3 After transition {OT} fires, as shown in Figure 3.15, the current output token of the gate overwrites the previous output token of the gate which was stored at place {P3}. The simulator time is 459. The output token at place {P4} is free to propagate to the Token Filter substructure of the HCAHPN. Formally, the values of tokens at all places are: P1 = (1, 100, P1, 459) P3 = (1, 0, P3, 459) P4 = (1, 0, P4, 459)

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51 3.2.3 Token Filter Substructure P4 P5 TF0 TF1 (v, p) (v, p) (v, p) (v, p) (v1, p1) (v1, p1) OP (v, p) [v != v1 | p1 = 100] [v = v1 & p1 != 100] place transition v, v1: Token Logic Variables p, p1: Token Port Number Variables Figure 3.16 Token Filter Structure Detail of AND Gate The details of the Token Filter Petri net substructure of a HCAHPN model for an AND gate are shown in Figure 3.16. This substructure is designed to limit simulation overhead by only propagating output tokens that are different from the previous output value. The exception to this rule is the very first output token. When a circuit first becomes operational, the state of a signal between two gates is unknown. Once the signal state between two gates is established, only changes in those signal states need to be propagated.

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52 Before the first output token propagates through the Token Filter substructure, the signal state of a wire between two gates is unknown. The visualization of this is shown in Figure 3.17. Once the first token passes through the Token Filter substructure and proceeds to the next substitution transition, the signal state between the two gates is known. The visualization of this is shown in Figure 3.18. Once the signal state between the two gates is known, only current output tokens that are different from the previous state of the output need to be propagated. X Figure 3.17 Unknown Signal State 1 Figure 3.18 Known Signal State Place {P5} is used to store the previous state of the output of the gate. Transitions {TF0, TF1} are used to control the propagation of output tokens. Transition {TF0} is enabled if the current value of the output matches the previous output value of the gate and the token present at place {P4} is not the very first output token propagating through the Token Filter substructure. Transition {TF1} is enabled if the current value of the output is different from the previous output value of the gate, or if the token present at place {P4} is the very first output token propagating through the Token Filter substructure. The enabling of transitions {TF0, TF1} are both mutually exclusive.

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53 There is no token present at places {P4, OP} when the Petri net is initialized. The initialization function initializes a token at place {P5} to the value (0, 100, P5, 0). 3.2.3.1 Token Filter Firing Example This section provides an example of how the Token Filter substructure of the AND gate HCAHPN evaluates tokens. P4 P5 TF0 TF1 (v, p) (v, p) (v, p) (v, p) (v1, p1) (v1, p1) OP (v, p) [v != v1 | p1 = 100] [v = v1 & p1 != 100] place transition v, v1: Token Logic Variables p, p1: Token Port Number Variables 0 100 Figure 3.19 Token Filter Step 0 In Figure 3.19, the Token Filter substructure has just been initialized. The current simulator time is 0. Formally, the value of a token at place {P5} is (0, 100, P5, 0).

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54 P4 P5 TF0 TF1 (v, p) (v, p) (v, p) (v, p) (v1, p1) (v1, p1) OP (v, p) [v != v1 | p1 = 100] [v = v1 & p1 != 100] place transition v, v1: Token Logic Variables p, p1: Token Port Number Variables 0 100 1 0 Figure 3.20 Token Filter Step 1 Continuing from the switching activity example, Figure 3.20 shows the arrival of the new token at place {P4}. Since this is the first token to propagate through the Token Filter structure, transition {TF0} is enabled regardless if the output value is a zero or one. After transition {TF0} fires, the current output token at place {P4} will overwrite the previous output token at place {P5}. The token is also sent to place {OP} where it can propagate through to different parts of the HCAHPN. The simulator time is 459. Formally, the values of tokens at all places are: P4 = (1, 0, P4, 459) P5 = (0, 100, P5, 0)

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55 P4 P5 TF0 TF1 (v, p) (v, p) (v, p) (v, p) (v1, p1) (v1, p1) OP (v, p) [v != v1 | p1 = 100] [v = v1 & p1 != 100] place transition v, v1: Token Logic Variables p, p1: Token Port Number Variables 1 0 1 0 Figure 3.21 Token Filter Step 2 Figure 3.21 shows the status of the Token Filter substructure just after transition {TF0} has fired. The token at place {P4} was consumed by the firing of the transition and deposited onto places {P5, OP}. The simulator time is 459. The token at place {OP} is free to propagate to other places. Formally, the values of tokens at all places are: P5 = (1, 0, P5, 459) OP = (1, 0, OP, 459)

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56 3.3 HCAHPN High-Level Modeling The goal of turning an asynchronous circuit into a HCAHPN is to accurately capture switching activity and the number of request and acknowledge signal assertions. During this conversion process, the circuit behavior and timing information must be preserved. As an example of this process, the wine shop [54] asynchronous circuit is shown in Figure 3.22. C C req_patron ack_patron req_wine ack_wine inv1_out inv2_outC1 C2 Figure 3.22 Wine Shop Asynchronous Circuit The wine shop asynchronous circuit consists of two inverters and two Muller C-elements. Muller C-elements are commonly used logic gates in asynchronous circuits and are usually not available in most synchronous standard cell libraries. Since this work integrates a synchronous standard cell library, Muller C-elements are realized by feeding a 3-input OR gate with three 2-input AND gates. The next step is to flatten the wine shop down to its primitive gate cells. The flattened wine shop can be seen in Figure 3.23.

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57 The flattened wine shop consists of tw o inverters, six AND gates, and two OR gates for a total of ten gates. Each of the gates are marked {G1, G2, …, G10}, and the switching activity to be captured at each gate output is marked {S1, S2, … S10}. It is important to note that gate {G4} is driving an acknowledge signal, and gate {G10} is driving a request signal. _ inv1_out inv2_outG1 G2 G3 G4G5G7 G8 G9 G10G6 S1 S2 S3 S4 S5 S6 S7 S8 S9 S10ack_patron ack_wine req_wineack_wine req_patron req_patron Figure 3.23 Flattened Wine Shop Circuit To obtain the HCAHPN structure for th e flattened wine shop, the flattened wine shop is first converted into its corresponding gate signal graph (GSG) as shown in Figure 3.24. Each gate of the circuit is represented as a node within the GSG, and each signal wire is represented as an arc between different nodes. The switching activity to be captured is also represented as a node within the GSG; additional arcs are introduced to maintain connectivity between the gate nodes. For gate outputs that have a fan-out greater than one, a node is introduced into the GSG to represent each fan-out signal. For example, the output signal {S5} from gate {G5} is driving two

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58 gates {G1, G3}. The node {S15} corresponds to the output signal from gate {G5} to gate {G1}, and the node {S14} corresponds to the output signal from gate {G5} to gate {G3}. To convert the GSG into a Petri net, signal nodes are converted into places, and gate nodes are converted into transitions. The equivalent Petri net for the flattened wine shop in Figure 3.23 is shown in Figure 3.25. req_wine1 G1 G2 G3 S1 S2 S3 G7 G8 G9 S7 S8 S9 G4 G10 G6 S6 S10 S12 S11 G5 S13 S14 S15 S17 S16 S18 S19 req_wine2 ack_patron Figure 3.24 Wine Shop Gate Signal Graph There are three different types of substitution transitions within a HCAHPN. Depending on the type of signal a gate is driving, its corresponding substitution transition will either be an ordinary, request, or acknowledge substitution transition. The switching activity recorded by subnets belonging to request and acknowledge substitution transitions also serves to record the number of request and acknowledge

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59 signal pairs. It is necessary to know the total number of request and acknowledge signal assertions in a circuit to calculate the total number of invocations. req_wine1 S1 S2 S3 S7 S8 S9 S6 S10 S12 S11 S13 S14 S15 S17 S16 S18 S19 req_wine2 ack_patron G1 G2 G3 G4 G6 G5 G7 G8 G9 G10 place transition Figure 3.25 Wine Shop Petri Net To transform the Petri net in Figure 3.25 into a HCAHPN, hierarchy, color, type, and delay information is added to the Petri net. Each transition corresponding to a gate is replaced by an appropriate substitution transition. As detailed in the previous section, each subnet represented by a substitution transition accurately models its corresponding gate. For circuit inputs that are driving multiple gates, each input is replaced by a transition and an appropriate number of places. For example, the input places {req_wine1, req_wine2} in the Petri net in Figure 3.25, are replaced

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60 by a single transition that is connected to two places. For gates that are driving request and acknowledge signal wires, they are replaced by their appropriate request and acknowledge substitution transitions. For example, since gate {G10} is driving a request signal, it is replaced by a corresponding request substitution transition. Interconnect delay is represented by adding delay information to arcs. The HCAHPN structure for the flattened wine shop is shown in Figure 3.26. req_wine S1 S2 S3 S7 S8 S9 S6 S10 S12 S11 S13 S14 S15 S17 S16 S18 S19 ack_patron place transition substitution transition request substitution transition acknowledge substitution transition G10 G9 G8 G7 G1 G2 G3 G3 G6 G5G4 T1 R1 R2 d d d d d d d d d d d d d d d d d d d d delay Figure 3.26 Wine Shop HCAHPN

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61 CHAPTER 4 A FRAMEWORK FOR ENERGY ESTIMATION In this chapter, a new framework for energy estimation of asynchronous circuits is described. The framework is shown in Figure 4.1. 4.1 Tool Flow Balsa Verilog Netlist Flattener Technology Conversion Script CPN Processor Xilinx Verilog Netlist VTVT Verilog Netlist Flattened Netlist Cadence Encounter Flattened Netlist DEF File SPEF File Interconnect Delay Calculator Layout Information Parasitics Gate Distances Interconnect Delay HSPICE Gate Models Intrinsic Gate Delays Primitive Gate Petri Net Models CPN Tools Simulator Input Vectors Simulation Energy Cadence PKS Cadence NCSim Toggle Information Flattened Netlist Energy Comparison Gate Capacitances Switching Activity Technology Library LEF / LIB files Technology Library LEF / LIB files XML File Figure 4.1 Proposed Framework for Energy Estimation The tool flow for this work is shown in Figure 4.1. A description of each of the tools used follows. Balsa is a high-level asynchronous synthesis tool. It is able to generate asynchronous circuits in a number of differe nt implementation styles, including fourphase handshaking and dual-rail delay insensitive. Balsa is produced and maintained

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62 by the Advanced Processor Technologies group at the University of Manchester in England. Balsa comes with support for Xilinx FPGA technology. For more on Balsa, the reader is referred to [49]. This work uses the Virginia Tech VLSI For Telecommunications VTVT TSMC 0.25um cell library [50, 51]. Instead of plugging the cell library directly into Balsa, a Perl conversion script was developed to map Xilinx gate primitives to VTVT standard cells. One of the features of netlists produced by Balsa is that high-level modules distinguish between request, acknowledge, and data wires. However, this signal information is lost as one progresses into each of the sub-modules in a Balsaproduced verilog netlist. In order to propagate high-level request, acknowledge, and data wire names down to the primitive gate level, a verilog netlist flattener was written. Preserving signal types to the final structural netlist is necessary to calculate the number of invocations in the circuit. Submodule One Submodule Two Request Main Module Acknowledge Data Main Module Request Acknowledge Data After Flattening Figure 4.2 Verilog Netlist Flattener Since this work uses a real delay model, and one must consider the effects of interconnect delay in deep submicron technologies like the TSMC 0.25um

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63 technology, an appropriate interconnect delay formula had to be chosen. This work implements the Elmore Delay for RLC Trees [52]. Other suitable interconnect models for deep submicron technologies include transmission line models [46] and transmission line models with coupling capacitances [47]. The interconnect delay [52] from an output node to a given node i on a signal line is given by: ni i pdie t / ) 39 1 047 1 () 85 0 / ( (4.1) k ik k k ik k iL C R C 2 1 (4.2) k ik k niL C 1 (4.3) where k is a value that covers all nodes of the RLC tree, Rik is the resistance from the input to nodes i and k Lik is the inductance from the input to nodes i and k, and Ck is the capacitive load at node k Cadence Encounter is used to place and route each flattened verilog netlist. Once the design has been placed and routed, the layout information can be extracted. From the Standard Parasitic Exchange Form at (SPEF) file generated by Encounter for each design, nodal capacitances and resistances can be extracted for each RLC tree. The native RC extractor in Encounter is una ble to extract inductances, however. To calculate interconnect inductances, the following formula[53] is used: l t w t w l l LO) ( 2235 0 2 1 2 ln 2 (4.4)

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64 where O is the permeability of free space, l is the length of the wire, w is the width of the wire, and t is the thickness of the wire. To compute the length of the wire, distance information for each net is extracted from the Design Exchange Format (DEF) file generated by Encounter for each design. The width and thickness of a wire is known from the technology file. The intrinsic gate delays for each standard cell used in the design were calculated using HSPICE. The delays were calculated assuming a standard capacitive load of a FO4 inverter, which in this case equals 0.02 picofarads. The table below shows the intrinsic gate delay for each standard cell utilized in picoseconds. Table 4.1 Intrinsic Gate Delays Intrinsic Gate Delays Cell Delay (ps) BUF1 132 INV1 110 NAND2114 NAND3146 NAND4190 AND2 159 AND3 201 AND4 263 NOR2 169 NOR3 248 NOR4 340 OR2 199 OR3 287 OR4 391 DRP 500 One assumption made in the implementation of HCAHPNs is that rise and fall transition times are the same. In determining the instrinic gate delay for each standard cell, both the rise and fall intrinsic gate delays were measured in HSPICE; the greater of the two numbers was chosen to represent the intrinsic gate delay. The

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65 gate delays incorporated into each HCAHPN primitive gate model represent the worst-case intrinsic gate delay. The CPN Processor is a software tool th at accepts as input a flattened verilog netlist, HCAHPN models for each primitive gate, and interconnect delay values for each net and produces as output a corresponding HCAHPN in XML format that can be simulated in CPN Tools. The CPN Proce ssor also generates test vector inputs for the CPN Tools simulator and the verilog test bench used by NCSim. The test vectors are randomly generated sequences of binary values. Since each circuit simulated is very small and has a limited number of inputs, a sequence of 10,000 input vectors for each circuit is more than adequate to guarantee complete test coverage. CPN Tools is a colored Petri net simulator developed by the CPN group at the University of Aarhus in Denmark. It includes the ability to simulate both timed and untimed colored Petri nets. With the ability to support hierarchical nets, complex systems can be abstracted and simulated as a Petri net in CPN Tools. The CPN Tools simulator accepts the Petri net generated by the CPN Processor as well as input stimulus files in order to simulate a circuit. After simulation of the Petri net is complete, the switching activity collected from the monitors in CPN Tools is known for each gate. The collected switching activity and load capacitance for each gate is used to calculate the amount of energy dissipated by each gate over the course of circuit simulation. As a reference point for the proposed energy estimation method, energy estimates for each circuit were obtained from the gate-level power estimator in Cadence PKS. The circuit was first simula ted in NCSim using the same input vectors

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66 as CPN Tools. The toggle information from simulation in NCSim is used to calculate the dynamic power consumption of each circuit in Cadence PKS. To obtain the dynamic energy consumption of the circuit, the total dynamic power consumption is divided by circuit simulation time. While Cadence PKS does not distinguish between asynchronous and synchronous circuits, this tool was integrated into the flow to show that existing tools for synchronous power estimation can be used on asynchronous designs. 4.2 Energy Estimation Formula This work estimates the average dynamic energy per invocation consumed in a circuit. The energy estimation formulas presented in [1] and [40] are combined to produce a new energy estimation formula that calculates the average dynamic energy consumed per invocation. The formula in [1] is: Einvocation = * 2 12dd loadVCD(transitions) + Pdt (4.5) where Vdd is the supply voltage, Cload is the capacitive load, D(transitions) is the transition density, and Pdt is the energy consumed by a delay line. The energy estimation formula used in [40] considers the energy consumed per output transition. The formula is: * 2 1 ) (2dd load gate gatesV T EnC (# of gate switches) (4.6) By combining the two previous formulas, it is possible to estimate the average dynamic energy consumed per invocation. This formula is: ) (# ) ( * 2 12s invocation switches V Egate dd load gate gates invocationC (4.7)

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67 where Cload gate is the capacitive output load of a gate, ddV is the supply voltage, gateswitches is the number of times the output of a gate has switched, and s invocation #is the total number of completed request/acknowledge handshake pairs.

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68 CHAPTER 5 EXPERIMENTAL RESULTS The methodology was tested for a total of eleven different circuits. Except for the wine shop circuit, all the circuits listed are four-phase handshaking implementations of high-level descriptions of asynchronous circuits produced from Balsa. Table 5.1 Simulation Results CPN Tools Simulation Cadence PKS Simulation Energy SimulationTotal Average EnergySimulationTotal Difference Name Circuit Description Gates Time (s) Energy (nJ) Invocations (E-4 nJ) Time (s) Energy (nJ) (%) shop Wine shop 10 70 0. 536 1247 4.298 12 0.406 32 buffer1a 1-bit buffer 36 142 1.222 10431 1.172 13 1.108 10 buffer1b 1-bit buffer 36 133 1.214 10345 1.174 13 1.105 10 buffer2a 2-bit buffer 41 162 1.502 10381 1.446 13 1.431 5 buffer2b 2-bit buffer 41 162 1.517 10669 1.422 13 1.439 5 demux1 2-bit demux 38 185 1.501 2293 6.546 12 1.250 20 demux2 3-bit demux 42 217 1.115 2506 4.449 14 1.554 39 merge1a 1-bit 2-input mux 56 229 2.097 12077 1.736 13 1.559 35 merge1b 1-bit 2-input mux 46 212 1.857 12491 1.487 14 1.510 23 merge2a 2-bit 2-input mux 64 279 2.418 12344 1.959 15 1.908 27 merge2b 2-bit 2-input mux 54 273 2.301 12747 1.805 15 1.831 26 The CPN Tool simulations were run on a Windows XP machine with an Athlon XP 3000+ with 1 gigabyte of RAM. The Cadence PKS simulations were run on a Sun OS 4 machine with dual 450 MHz pr ocessors with 2 gigabytes of RAM.

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69 During simulation in CPN Tools, the switching activity for each gate as well as the number of acknowledge and requests in the circuit are recorded. This activity is used to calculate the amount of energy dissipated by each gate as well as the number of total invocations in the circuit. For each circuit, a set of 10,000 random binary input vectors was generated. The sa me test vectors were used in CPN Tool simulation and Cadence PKS simulation. Fo r both simulation tools, the time between different input vectors was set to 500 ps. Th ere is a wide variation in the results for energy consumption between CPN Tools and Cadence PKS. While CPN Tools uses the actual capacitance from layout, Cadence PKS uses information from the technology library to estimate the load capacitance. These two different methods of calculating load capacitance can explain the wi de variation of results between the two simulators. One major limitation of the current version of CPN Tools is that it stalls on syntax checking on large numbers of monitors Since the monitoring functionality of CPN Tools is used to record the switc hing activity of each gate, the number of monitors in a simulation is a function of the number of gates within a circuit. The largest design CPN Tools was able to syntax check successfully was merge2a, which is 64 gates; the simulator stalled on syntax checking of a 74 gate two-bit asynchronous shifter. Without monitoring functionality present, CPN Tools was able to complete the loading of Petri nets with several hundred gates. Once this limitation is corrected in future versions of CPN T ools, the methodology should be able to work with designs ranging from hundreds to thousands of gates.

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70 While interconnect delay is considered in the simulation of the circuits, the delay effects are negligible due to the very small circuit sizes. Once the major limitation of CPN Tools is corrected, the effects of interconnect delay would be clearly seen as circuits scale into the hundreds and thousands of gates. In this thesis, the objective was to develop a framework for Petri net modeling of asynchronous circuits for energy estimation. Thus, we have developed new models for asynchronous gates and circuits, developed a complete framework and tool flow for energy estimation, and implem ented the proposed tool using CPN Tools. The use of CPN Tools is only to show proof of concept by implementing a customized software tool based on our proposed methodology with results in a time efficient tool that can handle large circuits. A practical tool needs to be built in the future for use in asynchronous circuit design environment.

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71 CHAPTER 6 CONCLUSIONS This work is the first to do simulative gate-level energy estimation of asynchronous circuits. This work also includes a real-delay model to capture the effects of glitch power. While interconnect delay was negligible next to gate delay in the very small circuits that were tested, once larger circuits with hundreds and thousands of gates can be tested, one would clearly see the effects of interconnect delay and glitch power. Once the limitation in CPN Tools has been corrected, larger designs of several hundred and even thousands of gates could be simulated. While test vector generation for exhaustive simulation of a circuit is not a problem given the very small circuit sizes that were simulated, it does become a problem with designs that range in the hundreds and thousands of gates. Future work could include state space exploration to produce appropriate test vectors. CPN Tools has been designed for many us es. Many different types of systems can be modeled as HCPNs, which CPN Tools has the capability to simulate. In its current state, however, CPN Tools is inefficient for hardware simulation. The proposed methodology is only suitable for asynchronous circuit implementations that use handshaking communication on distinct request and

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72 acknowledge signal lines. For timed asynchronous circuit implementations that combine request and acknowledge channels into one signal wire, the proposed methodology is not suitable for energy estimation.

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73 REFERENCES 1. P. Kudva and V. Akella, “A Technique for Estimating Power in Asynchronous Circuits”, Proceedings of the International Symposium on Advanced Research in Asychronous Circuits and Systems pp. 166-175, November 1994. 2. P. Gupta and A. B. Kahng, “Quantifyi ng Error in Dynamic Power Estimation of CMOS Circuits”, Proceedings on the Fourth International Symposium on Quality Electronic Design pp. 273-278, March 2003. 3. A. K. Murugavel and N. Ranganathan, “Petri net Modeling of Gate and Interconnect Delays for Power Estimation”, IEEE Transactions on Very Large Scale Integration (VLSI) Systems pp. 921-927, October 2003. 4. A. K. Murugavel and N. Ranganathan, “Power Estimation of Sequential Circuits using Hierarchical Colored Hardware Petri net Modeling”, Proceedings of the 2002 International Symposium on Lo w Power Electronics and Design pp. 267 – 270, 2002. 5. A. K. Murugavel and N. Ranganathan, “A Real Delay Switching Activity Simulator based on Petri net Modeling”, Proceedings of the 7th Asia and South Pacific and the 15th International Conference on VLSI Design pp. 181-186, Jan 2002. 6. C.D. Bagwell, E. Jovanov, and J. H. Julick, “A Dynamic Power Profiling of Embedded Computer Systems”, Proceedings of the Thirty-Fourth Southeastern Symposium on System Theory pp. 15-19, March 2002. 7. C. Talarico, J. W. Rozenblit, V. Malhotra, and A. Stritter, “A New Framework for Power Estimation of Embedded Systems”, Computer Vol. 38, No. 2, pp. 71-78, February 2005. 8. Z. Huang and P. Zhong, “An Architectur al Power Estimator for Analog-to-Digital Converters”, Proceedings of the IEEE International Conference on Computer Design: VLSI in Computers and Processors, pp. 397-400, October 2004. 9. T. Givargis, F. Vahid, and J. Henke l, “Instruction-Based System-Level Power Evaluation of System-on-a-Chip Peripheral Cores”, IEEE Transactions on Very Large Scale (VLSI) Systems Vol. 10, No. 6, pp. 856-863, December 2002.

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74 10. R. Ludewig, A.G. Ortiz, T. Murgan, a nd M. Glesner, “Power Estimation based on Transistion Activity Analysis with an Architecture Precise Rapid Prototyping System”, Proceedings of the 13th IEEE International Workshop on Rapid System Prototyping pp. 138-143, July 2002. 11. C. Isci and M. Martonsi, “Runtime Power Monitoring in High-End Processors: Methodology and Empirical Data”, Proceedsings of the 36th Annual IEEE/ACM International Symposium on Microarchitecture pp. 93-104, 2003. 12. K. Chang, C. Weng, and S. Huang, “Accurate RTL power estimation for a security processor”, Conference on Emerging Information Technology pp. 89-91, August 2005. 13. M. Sum, S. Huang, C. Weng, and K. Chang, “Accurate RT-level power estimation using up-down encoding”, Proceedings of the 2004 IEEE Asia-Pacific Conference on Circuits and Systems, pp. 69-72, December 2004. 14. D. Bruni, G. Olivieri, A. Bogliolo, and L. Benni, “Delay-Sensitive Power Estimation at the Register-Transfer Level”, The 8th IEEE International Conference on Electronics, Circuits and Systems pp. 1031 – 1034, September 2001. 15. F. Machado, Y. Torroja, and T. Riesgo, “Exploiting VHDL-RTL Features to Reduce the Complexity of Power Estimation in Combinational Circuits”, PhD Research in Microelectronics and Electronics pp. 111-114, July 2005. 16. M. Eirmann and W. Stechele, “N ovel modeling techniques for RTL power estimation”, Proceedings of the 2002 International Symposium on Low Power Electronics and Design pp. 323-328, 2002. 17. J. Coburn, S. Ravi, and A. Raghunathan “Power Emulation: A New Paradigm for Power Estimation”, Proceedings of the 42nd Design Automation Conference pp. 700705, June 2005. 18. D. Tran, K.K. Kim, and Y. Kim, “Power Estimation in Digital CMOS VLSI Chips”, Proceedings of the IEEE Instrume ntation and Measurement Technology Conference pp. 317-321, May 2005. 19. C. Chen and M. Ahmadi, “Register-transfer-level power estimation based on technology decomposition”, IEE Proceedings on Circuits, Devices and Systems Vol. 150, No. 5, pp. 411-415, October 2003. 20. M. Eirmann and W. Stechele, “RTL Power Modeling and Estimation based on bit and word level Switching Properties”, The 2002 45th Midwest Symposium on Circuits and Systems Vol. 1, pp. 144-147, August 2002.

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75 21. J.L. Rossello, C. de Benito, and J. Segura, “A Compact Gate-Level Energy and Delay Model of Dynamic CMOS Gates”, IEEE Transactions on Circuits and Systems II: Express Briefs Vol. 52, No. 10, pp. 685-689, October 2005. 22. W. Shiue, “Accurate Power Estimation for CMOS Circuits”, Proceedings of IEEE Region 10 International Conference on Electrical and Electronic Technology Vol. 2, pp. 829-833, August 2001. 23. M. Tahrezadeh-S, B. Amelifard, H. Iman-Eini, F. Farbiz, A. Afzali-Kusha, and M. Nourani, “Power and Delay Estimation of CMOS Inverters using Fully Analytical Approach”, Southwest Symposium on Mixed-Signal Design pp. 116-120, February 2003. 24. S. Jung, J. Baek, and S. Kim, “Short Circuit Power Estimation of Static CMOS Circuits”, Proceedings of the ASP-DAC 2001 Asia and South Pacific Design Automation Conference pp. 545-549, February 2001. 25. S.S. Ramani and S. Bhanja, “Any-time Probablistic Switching Model using Bayesian Networks”, Proceedings of the 2004 International Symposium on Low Power Electronics and Design pp. 86-89, 2004. 26. S. Bhanja and N. Ranganathan, “Cascaded Bayesian inferencing for Switching Activity Estimation with Correlated Inputs”, IEEE Transactions on Very Large Scale Integration (VLSI) Systems Vol. 12, No. 12, pp. 1360-1370, December 2004. 27. L. Cao, “Circuit Power Estimation using Pattern Recognition Techniques”, IEEE/ACM International Confer ence on Computer Aided Design pp. 412-417, November 2002. 28. A. Dogandzic and B. Zhang, “Dynamic Power Estimation and Prediction in Composite Fading-Shadowing Channels”, Proceedings of the IEEE International Conference on Acoustics, Speech, and Signal Processing Vol. 2, pp. 1013-1016, May 2004. 29. R. L. Wright and M. A. Shanblatt, “Improved Power Estimation for Behavioral and Gate Level Designs”, Proceedings of IEEE Computer Society Workshop on VLSI pp. 102-107, April 2001. 30. S. Bhanja and N. Ranganathan, “Switching Activity Estimation of VLSI Circuits using Bayesian Networks”, IEEE Transactions on Very Large Scale Integration (VLSI) Systems Vol. 11, No. 4, pp. 558-567, August 2003. 31. X. Liu and M.C. Papaefthymiou, “A Markov Chain Sequence Generator for Power Macromodeling”, IEEE/ACM International Conf erence on Computer Aided Design pp. 404-411, November 2002.

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76 32. N.E. Evmorfopoulos, G.I. Stamoulis, and J.N. Avaritsiotis, “A Monte Carlo Approach for Maximum Power Estimation based on Extreme Value Theory”, IEEE Transactions on Computer-Aided Design of Integrated Circuits and Systems Vol. 21, No. 4, pp. 415-432, April 2002. 33. X. Liu and M.C. Papaefthymiou, “A St atistical Model of Input Glitch Propagation and its Application in Power Macromodeling”, The 2002 45th Midwest Symposium on Circuits and Systems Vol. 1, pp. 380-383, August 2002. 34. A.K. Murugavel, N. Ranganathan, R. Chandramouli, and S. Chavali, “Average Power in Digital CMOS Circuits using Least Square Estimation”, Fourteenth International Conference on VLSI Design pp. 215-220, January 2001. 35. A.T. Freitas and A.L. Oliveira, “Implicit resolution of the Chapman-Kolmogorov Equations for Sequential Circuits: an Application in Power Estimation”, Design, Automation and Test in Europe Conference and Exhibition pp. 764-769, 2003. 36. J. N. Kozhaya and F. N. Najm, “Power Estimation for Large Sequential Circuits”, IEEE Transactions on Very Large Scale Integration (VLSI) Systems Vol. 9, No. 2, pp. 400-407, April 2001. 37. A. G. Ortiz, L.D. Kabulepa, and M. Glener, “Transition Activity Estimation for Generic Data Distributions [Power Estimation]”, IEEE International Symposium on Circuits and Systems Vol. 2, pp. 468-471, May 2002. 38. K. Jensen, Colored Petri nets : Basic Concepts 2nd ed., Berlin, Germany: Springer-Verlag, Vol. 1, 1996. 39. L. Lloyd, A.V. Yakovlev, E. Pastor, and A. M. Koelmans, “Estimations of Power Consumption in Asynchronous Logic as Derived from Graph Based Circuit Representations”, Power and Timing Modeling, Optimization and Simulation (PATMOS) 1998. 40. P. A. Beerel, K. Y. Yun, S. M. No wick, and P.-C. Yeh, “Estimation and bounding of energy consumption in burst-mode control circuits”, 1995 IEEE/ACM International Conference on Computer-Aided Design pp. 26-33, November 1995. 41. N. Weste and D. Harris, CMOS VLSI Design: A Circuits and Systems Perspective 3rd ed., Pearson Education, 2005. 42. C. Tsui, M. Pedram, and A. M. Despai n, “Efficient Estimation of Dynamic Power Consumption under a Real Delay Model”, 1993 IEEE/ACM International Conference on Computer-Aided Design pp. 224-228, November 1993.

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77 43. T. Chou and K. Roy, “Estimation of Activity for Static and Domino CMOS Circuits Considering Signal Correlations and Simultaneous Switching”, IEEE Transactions on Computer-Aided Design of Integrated Circuits and Systems Vol. 15, No. 10, pp. 1257-1265, October 1996. 44. K.M. Buyuksahin and F.N. Najm, “High-level power estimation with interconnect effects”, Proceedings of the 2000 International Symposium on Low Power Electronics and Design pp. 197-202, 2000. 45. R. Marculescu, D. Marculescu, and M. Pedram, “Switching Activity Analysis Considering Spatiotemporal Correlations”, 1994 IEEE/ACM International Conference on Computer-Aided Design pp. 294-299, November 1994. 46. T.-C. Chen, S.-R. Pan, and Y.-W. Chang, “Timing Modeling and Optimization under the Transmission Line Model,” IEEE Transactions on VLSI Systems vol. 12, no. 1, pp. 28-41, January 2004. 47. N. Hanchate and N. Ranganathan, “Simultaneous Interconnect Delay and Crosstalk Noise Optimization through Gate Sizing Using Game Theory”, IEEE Transactions on Computers vol. 55, No. 8, pp. 1011-1023, August 2006. 48. P.A. Beerel, C. Hsieh, and S. Wadekar, “Estimation of Energy Consumption in Speed-Independent Control Circuits,” IEEE Transactions on Computer-Aided Design of Integrated Circuits and Systems vol. 15, No. 6, pp. 672-680, June 1996. 49. A. Bardsley. The Balsa web pages. http://intranet.cs.man.ac.uk/ apt/projects/tools/balsa/. 50. J. B. Sulistyo, J. Perry, and D. S. Ha, Developing Standard Cells for TSMC 0.25um Technology under MOSIS DEEP Rules", Department of Electrical and Computer Engineering, Virginia Tech, Technical Report VISC2003-01 November 2003. 51. J. B. Sulistyo and D. S. Ha, "A New Characterizati on Method for Delay and Power Dissipation of St andard Library Cells", VLSI Design, Vol. 15, No. 3, pp. 667-678, 2002. 52. Y. I. Ismail, E.G. Friedman, and J. L. Neves, “Equivalent Elmore delay for RLC trees,” IEEE Transactions on Computer-Aided Design of Integrated Circuits and Systems vol. 19, No. 1, pp. 83-97, January 2000. 53. F. Grover, Inductance Calculations Working Formulas and Tables Instrum. Soc. Of America, 1945.

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78 54. C. J. Myers, Asynchronous Circuit Design New York: John Wiley & Sons Inc., 2001. 55. K. Jensen. “An introduction to the theoretical aspects of colored petri nets.” Advances in Petri nets Ed. J.W. Bakker; W.-P. de Roever; G. Rozenberg, LCNS 803 (1993), pp. 203-272, 1993. 56. International Road-map Committee. International Technology Road-map for Semiconductors 2006 update edition, 2006. 57. D. McFarland, Cell Phone Ownership Grows 29 Percent from 1999-2001, Scarborough Research. 18 March 2002. 58. S.B. Furber, J.D. Garside, P. Riocreux, S. Temple, P. Day, L. Jianwei, and N.C. Paver, “AMULET2e: an asynchronous embedded controller,” Proceedings of the Third International Symposium on Advanced Research in Asynchronous Circuits and Systems, 1997 pp. 290-299, 7-10 April 1997. 59. S. Rotem, K. Stevens, R. Ginosar, P. B eerel, C. Myers, K. Yun, R. Kol, C. Dike, M. Roncken, and B. Agapiev, “RAPPI D: an asynchronous instruction length decoder,” Proceedings of the Fifth International Symposium on Advanced Research in Asynchronous Circuits and Systems, 1999 pp.60-79, 1999. 60. P.L. Penzes and A. J. Martin, “An energy estimation method for asynchronous circuits with application to an asynchronous microprocessor,” Proceedings of the 2002 Design, Automation, and Test in Europe Conference and Exhibition pp. 640-647, 4-8 March 2002. 61. M. Salehi, K. Saleh, H. Kalantari, M. Naderi, and H. Pedram, “High-level energy estimation of template-based QDI asynchronous circuits based on transition counting,” ICM 2004 Proceedings of the 16th International Conference on Microelectronics, 2004 pp. 489-492, 6-8 Dec. 2004. 62. M. Niknahad, B. Ghavami, M. Najibi, and H. Pedram, “A Power Estimation Methodology for QDI Asynchronous Circuits based on High-Level Simulation,” IEEE Computer Society Annual Symposium on VLSI, 2007 pp. 471-472, March 2007.